One important field of application for the OFDM transmission method is high-rate, wire-free data transmission networks such as WLAN (Wireless Local Area Network), in particular the transmission methods defined in the standards IEEE 802.11a/g and ETSI TS 101 761.1 (BRAN, Broadband Radio Access Networks). The OFDM transmission method is a multicarrier transmission method in which the data stream is subdivided between a number of parallel (orthogonal) subcarriers, which are each modulated at a correspondingly low data rate. K (sub) carrier frequencies are arranged at equidistant intervals from one another within a transmission bandwidth on the frequency scale. The carrier frequencies are located on both sides of and symmetrically with respect to a mid-frequency fc. In the time domain, this results in an OFDM symbol produced by superimposition of all K carrier frequencies. The data is transmitted in the form of frames or bursts, with a frame having a structure which is defined by the standard.
FIG. 1 shows a section of a data burst which is transmitted from the transmitter end in accordance with the WLAN Standard IEEE 802.11a/g, and which starts at a time on the left-hand end, continues on the time axis to the right, and whose illustration is terminated after the second data symbol at the right-hand end. The data burst has a so-called PLCP preamble (Physical Layer Convergence Protocol), which is known from IEEE Standard 802.11a/g. This Standard describes the OFDM transmission method as a multicarrier transmission method. The payload data starts with the first OFDM symbol, which is denoted by “Data 1”. Each of the OFDM symbols has a length of 3.2 μs, and each OFDM symbol is preceded by a guard interval (GI) which corresponds approximately to the maximum expected duration of the impulse response of the transmission channel (in this case 0.8 μS).
The PLCP preamble has a length of four OFDM symbols, thus lasting for a total of 16 μs, and is subdivided into two sections, which each have the same time duration. A first section of the PLCP preamble with a length of 8 μs is subdivided into ten short symbols (also referred to in the following text as short training sequences or sections), which are used for signal detection, for automatic level matching, for diversity selection, for coarse frequency determination and for time synchronization in the receiver. The second section of the PLCP preamble comprises a G12 guard interval (of twice the length of the GI) and two OFDM symbols T1 and T2 (which are also referred to in the following text as long training sequences or sections). These data symbols are used for channel estimation in the receiver.
The reception and demodulation of OFDM radio signals can be carried out by conventional reception concepts, which are based on the principle of superheterodyne reception followed by digital quadrature mixing. However, particularly for reasons relating to reduced power consumption and avoidance of chip-external filters for mirror-image frequency suppression, more advanced reception concepts are increasingly being preferred, which use direct-mixing methods. In the case of direct-mixing receiver concepts, the radio signal which has been received via an antenna and has been amplified is split into an in-phase (I) path and a quadrature (Q) path, and is mixed with the output frequency from a local oscillator in both paths, with the oscillator frequencies which are supplied to the mixers being shifted through 90° with respect to one another by a phase shifter. The quadrature demodulation process for recovery of the baseband signals which contain the information is thus carried out using analogue circuit technology in this reception concept.
However, it is known that direct-mixing receiver structures have a tendency to add a DC voltage [DC] offset to the received signal, for various reasons. The negative influence of this DC offset is illustrated with reference to FIG. 2. The OFDM signal may be regarded as a superimposition of N modulated carrier frequencies. The frequency separation between the carriers is constant, and the signal is generated by Fourier transformation. In the WLAN application, the carrier with the index 0 is not used, and a DC offset could thus be tolerated. In order to allow the signal to be decoded, each carrier is filtered in the receiver with a filter whose filter function is a sinc function (f(x)=sin(x)/x), whose centre is at the carrier frequency. The zero points of the filter correspond to the position of the adjacent carrier frequencies, so that none of the carriers interfere with any of the other carriers. This situation in fact occurs only when there is no carrier frequency offset between the transmitter and the receiver, or between the carrier frequency of the receiver signal and the carrier frequency selected in the receiver. This situation is illustrated in FIG. 2A, in which the DC offset is the carrier placed at the frequency zero. Thus, in this specific case, the filter has a zero point exactly at the DC offset, so that there is no negative effect on the signal.
FIG. 2B illustrates the situation which always occurs in reality, in which there is also a carrier frequency offset. The DC offset is once again at the frequency zero, but the filter no longer has a zero point at this position. Some of the energy in the DC offset thus produces additional noise at the carrier frequencies. All of the carriers are thus adversely affected by this additional noise. The noise has a greater effect for carriers that are closer to DC, where the filters have a greater pass capability, than for carriers that are further away.
This has a negative influence in two ways on the signal decoding:
1) The channel estimation process, which is carried out with the long training symbols T1 and T2, is adversely affected relatively severely by the noise. The results of this channel estimation process are then used for equalization of all the subsequent OFDM data symbols.
2) The OFDM data symbols with the additional noise on them are equalized.
One method which is known from the prior art for estimation of the DC offset and for subsequent correction of or compensation for it is based on the fact that the WLAN signal is equal to zero on average, so that accumulation over an integer number of cycles of a periodic signal should produce a result which is proportional to the DC offset. In the prior art, this was done on the basis of a section of the ten short training symbols at the start of the PLCP preamble. In the following text, this section is also referred to as the short training sequence, since a sequence of digitized values is formed from each training symbol by A/D conversion.
FIG. 3 shows the signals that occur in the short training sequence, in the time and frequency domains. Twelve of the 52 carriers are modulated in this training sequence (FIG. 3B). The resultant signal has a periodicity of 0.8 μs (16 samples at 20 MHz). The 0.8 μs sequence is repeated ten times in the transmitter in order to produce the short training sequence as shown in FIG. 1. Since no DC carrier is used, the resultant 0.8 μs sequence has a mean value equal to zero. FIG. 3A shows the signal waveforms of the I and Q components. If one attempts to average one of these signals over a time interval which corresponds to a multiple of 0.8 μs, then this should result in the value zero irrespective of the phase angle at which the accumulation process was started. If a DC offset is in consequence superimposed on the sequence, then the result of the averaging process should give the DC offset itself.
FIG. 4 illustrates a situation in which three short sequences (3×16=48 samples at 20 MHz) have been averaged. The averaging process has a frequency-dependent pass capability in the frequency domain corresponding to the filter curve illustrated in FIG. 4. The intensity of the frequencies that are filtered by the averaging process is thus plotted on the ordinate of this diagram. The diagram additionally shows the twelve carriers which are modulated in the short training sequence and which are located exactly at the zero points of the frequency response of the filter function only if there is no carrier frequency offset, so that, in this unrealistic situation, they do not adversely affect the estimation of the DC offset.
FIG. 5 shows the same situation for the realistic case in which there is a carrier frequency offset. The maximum permissible frequency shift is ±250 kHz, with FIG. 5 showing the situation for a carrier frequency offset of −250 kHz. In this situation, the pilot carriers of the short training sequence are no longer located at the zero points of the filter function, and thus form a noise contribution to the averaging process. The averaging process was in this case carried out over an interval of 2.4 μs.
As is shown in FIG. 5, the energy in each attenuated carrier can be added up in order to estimate the noise power generated during the accumulation process, which is in the order of magnitude of −25 dB compared to the signal strength. The thermal noise should also be added to this noise contribution, whose noise power is in the order of magnitude of −10×log 10 (3×16)=−16.8 dB compared to the input noise power.
This situation is admittedly improved if more than three short sequences of the short training sequence are accumulated. However, the maximum number of sequences which are available for the estimation process is restricted, since the short training sequence comprises only ten sequences. Depending on the architecture of the reception path, some of the first sequences may be lost (owing to burst detection, gain adjustment, transients, etc.), and the number of usable, correct sequences may thus be further restricted. It is worth constraining this total number to be between two and five short sequences.
Another strategy for DC offset estimation is to carry out the estimation process during the long training sequence, which comprises the two OFDM symbols T1 and T2 (or along the OFDM data symbols which contain the payload data), to store this sequence and to correct it for the DC offset when the DC offset estimation is available, and not to use it for the channel estimation process until this has been done. In this situation, DC offset estimation over 128 samples would be possible. This results in a noise contribution to the estimation process resulting from the thermal noise of about −10×log 10(128)=−21 dB compared with the input noise power. In this case, the problem lies in the signal itself. All 52 carriers are modulated in this part of the preamble (and in all subsequent OFDM symbols). FIG. 6 shows the situation with a carrier frequency offset of −240 kHz, with the averaging process having been carried out over a section of the long training sequence of 6.4 μs. A coarse estimate of the noise that is produced by the signal itself is about −20 dB of the signal power, that is to say 5 dB worse than the noise contribution in the case of the short training sequence. In order to obtain noise contribution values which are in the same order of magnitude as the noise values obtained for the DC offset estimation process based on the short training sequence, the accumulation process must be carried out over at least four OFDM symbols. It is thus impossible to obtain a reliable DC offset estimate before the channel estimation process.